Sincap kafesli asenkron makinanın kayma kipli vektör kontrolü
Başlık çevirisi mevcut değil.
- Tez No: 55893
- Danışmanlar: DOÇ.DR. İBRAHİM EKSİN
- Tez Türü: Yüksek Lisans
- Konular: Bilgisayar Mühendisliği Bilimleri-Bilgisayar ve Kontrol, Computer Engineering and Computer Science and Control
- Anahtar Kelimeler: Belirtilmemiş.
- Yıl: 1996
- Dil: Türkçe
- Üniversite: İstanbul Teknik Üniversitesi
- Enstitü: Fen Bilimleri Enstitüsü
- Ana Bilim Dalı: Belirtilmemiş.
- Bilim Dalı: Belirtilmemiş.
- Sayfa Sayısı: 66
Özet
ÖZET Bu yüksek lisans tezi çalışmasında, sabit doğru gerilim aradevreli frekans çe virici ile vektör kontrol yöntemiyle sürülen sincap kafesli asenkron makinanm kayma kipli hız kontrolü gerçekleştirilmiştir. Çalışmada hedeflenen temel amaç, belirsizlikler içeren ve çalışma parametreleri değişen sincap kafesli asenkron makinanm yüksek performanslı kontrolüdür. Sincap kafesli asenkron makinanm yüksek performanslı kontrolü hem sürek li hem de geçici hal rejiminin kontrol edilebildiği vektör kontrol teknikleri ile yapılabilir. Vektör kontrolde temel amaç, elektrik ve mekanik kısımları arasında nonlineer kuplaj bulunan ve bu yüzden büyüklükleri ayrı ayrı kontrol edilemeyen asenk ron makinanm, dekuple edici bir kontrolla doğru akım makinasma benzer hale getirilmesidir. Hatırlanacağı üzere doğru akım makinasmda moment uygulanan akımla lineer olarak değişmektedir. Vektör kontrol uygulanmasıyla asenkron makinada akı ile moment arasındaki kuplaj ortadan kalkmakta ve bu iki büyüklük ayrı ayrı kontrol edilebilmektedir. Makina parametrelerinde, özellikle çalışma esnasında makina yük alıp yük atar ken meydana gelen yük momentindeki değişimlere karşı dayanıklı kontrol sis temleri gerçekleştirmek için, yaygın biçimde kullanılan nonlineer kontrol yön temlerinden biri olan, değişken yapılı kontrol prensipleri kullanılmıştır. Çalışmada, değişken yapılı kontrol yöntemlerinden biri olan kayma kipli kontrol kullanılmıştır. Bu yöntemin uygulanmasıyla asenkron makinanm hız kontrolü için gerekli kontrolör dizaynları bu yöntemle yapılmıştır. Kayma kipli kontrolde karşılaşılan ve çıtırtı olarak adlandırılan anahtarlama sonucu oluşan hareket de süreksiz kontrolün belirli bölgelerde süreklilendirilme- siyle azaltılmıştır. Sincap kafesli asenkron makinaya uygulanan kayma kipli vektör kontrol yönte minin performansı yapılan bilgisayar simulasyonları ile incelenmiş, karşılaştırıl mış ve sonuçlar tatminkar bulunmuştur. vııı
Özet (Çeviri)
SUMMARY VARIABLE STRUCTURE CONTROL OF VECTOR CONTROLLED INDUCTION MACHINE In this work, the variable structure speed control of vector controlled squirrel cage induction machine is investigated. It is estimated that 75% of all electric motor driven applications require the speed of the motor to be reduced, or the torque to be increased or both. The bulk of these applications concerns simple geared units or variable speed fans, compressor and pump type loads. High performance servo applications are on the increase, primarily driven by the substantial growth in manufacturing automation. Increasing numbers of robots and other forms of custom automation meet the ever-increasing needs for improved product quality and greater productivity. Motion control is the basic underlying requirement for almost all of this au tomation. Motion control can be defined as the application of high performance servo drives to rotational or translational control of torque, speed, and/or position. At present, numerous systems are available for such purposes including dc motor drives, variable reluctance stepper drives, and brushless dc motor drives. The greatest progress has perhaps been made in recent years by the induction motor servo thanks to field oriented control which allowed the induction motor to move beyond the variable speed control of Volt/Hertz drives. Induction motors have been widely applied in the motion control because of their very wide speed range, ruggedness, low maintenance, and relatively low cost. Advances in power electronics and microprocessor technology make it feasible to use the induction motors in place of dc and synchronous motors in a wide range of servo applications. However, it is very difficult to achieve high performance with an induction motor due to the intrinsic nonlinear coupling between the dynamics of the electrical part and of the mechanical (torque) part. In a seperately excited dc motor, the electromechanical arrangement avoids the dynamics coupling between the stator side and the rotor side, apparent in the torque dynamical equation. The dc motor can be linearly operated simply by keeping the stator current constant. For an induction motor, however, the stator windings will generate a flux which indirectly creates a rotor flux as well. Angular separation between the two fluxes results from the time delay inherent in the rotor circuit. Since the stator current here is the only source of generating torque, the stator and rotor effects ixwill obviously have to be tightly coupled. This explains why the induction machine cannot be operated as a linear device. The speed control of the induction motor has a paramount importance for industrial applications. The speed control of induction motors can be divided into two distinct categories depending on the type of dynamics required: i) Scalar control ii) Vector control (or field oriented control) Scalar control involves maintaining the flux magnitude in the machine constants. The Volt/Hertz controllers which use voltage source inverters to obtain variable voltage-variable frequency control by pulse width modulation strategies, fall under this category. Though this control method has the advantage of simplicity and low cost, its main drawback is the poor and sluggish torque dynamics. In the case of vector controllers, the current space vector is controlled both in magnitude and position to achieve decoupled control of the torque producing and the flux producing components of the stator current space phasor. This allows good transient response of the motor. To achieve decoupled control, either the stator flux, airgap flux or the rotor flux should be known both in magnitude and position. This is usually obtained by using either flux sensors (direct field oriented control) or by estimators using measurable states of the induction motor (indirect field oriented control). The implementation of the direct field oriented control requires the measure ment or calculation of the flux space phasor both in magnitude and position. Sensing devices placed in the airgap of the machine will determine the airgap flux which can be sensed by using either Hall effect devices or stator search (sense) coils. But both techniques suffer from the common disadvantage that a specially constructed induction motor is required. Hall sensors are very sensitive to temperature and mechanical vibration, and also the flux signal is distorted by large slot harmonics that cannot be filtered effectively because their frequency varies with motor speed. In the case of stator search coils that sense the rate of change of airgap flux, induced voltage in the search coils is proportional to the rate of change of flux. At low speeds below about 1 Hz, the induced voltage will be significantly low, it will give rise to inaccurate flux sensing owing to the presence of noise and disturbances in a practical system. On the other hand, indirect sensing of flux space phasors gives a more versatile drive system that can be used with standard commercial motors, but this ap proach would generally result in a more complex control system. In the indirect method of field orientation, the flux space phasor is estimated from the motor model. As a consequence, all indirect methods are sensitive to variations insome machine parameter. Figure 3.3 explains the indirect vector control principle with help of a phasor diagram. The ds - qs axes are fixed on the stator while the de - qe axes rotate at synchronous angular velocity uje as shown. At any instant, the qe electrical axis is at angular position 9e with respect to the qa axis. The angle 9e is given by the sum of rotor angular position 9m and slip angular position 0r, where 8e = wei, 9m = wmt, and 9r = ujrt. The rotor flux i/>r, consisting of the air gap flux and the rotor leakage flux, is aligned to the de axis as shown. Therefore, for decoupling control, the stator flux component of current ids and the torque component of current iqs are to be aligned to the de and qe axes, respectively. We can write the following equations for rotor flux components + Rriqr + («e ~ Fm)^r = 0 + Rridr - (Wc - pum)ll)qr = 0 dlpqr dt dlpdr dt and we have the following flux equations Ipqr = ?Lir^qr T J-'m'^qs lf)dr = Lrİdr + Lmİda or solving these equations for iqr and idr we have Iqr - ~~f Wqr ? j Iqs - - I m. ^dr - T Wdr T Ids The rotor currents from differential equations above can be eliminated by sub stituting these last two equations as dipgr R dt ' Lr''“11 L + -l]>qr ^Rrİqs + ^rİ^dr = 0 jr ±jt dlpdr. Rr i Lm in + -ydr -Rrlds ~ e - pum. For decoupling control it is desirable that Ipqr =0 = 0 dxj)qr dt İ>dr = Vv = constant dlpdr dt r 0 XISubstituting the first two conditions, we can obtain -”m f ?tw - 1pr Lr Lr dlf>r Yr~dT + ^ = Lmtds The torque as a function of rotor flux and stator current can be given as follows: Substituting ipgr = 0 and tpdr = Vv> the torque expression is 3 Lm 2i LdT Te = -p-^iqS^r The equations above, together with the mechanical equation _ÛWm _ _ describe the machine model in decoupling control as shown in Figure 3.4. The inverter is assumed to be current controlled, and the delay between the command and response currents are neglected. The developed torque Te responds instantaneously with current iqs, but has delayed response due to ids- The analogy of the model with a seperately excited dc machine is obvious. Application of a variable structure control strategy using sliding mode has re ceived worldwide acceptance in the recent past for controlling electric drive sys tems. Sliding mode control offers a parameter sensitive control feature, a fast dynamical response, a variable control system, and a minimal implementation hardware. Most high performance ac drives incorporate inner-loop vector controllers as discussed above. The complex nonlinear characteristics of the ac motors are decoupled (torque and flux) by these inner-loop controllers. Conventionally, a PI controller is used in the outer loop for the compensation of drive operating characteristics in the presence of external load and param eter disturbances. However, it is well known that the performance obtained by PI controllers is sensitive to plant parameter variations. Furthermore, the PI controller gains have to be carefully selected in order to obtain a desired response. The advent of microelectronics and microprocessors has provided a strong incentive for the applications of nonlinear control theories in order to obtain improved drive responses and stability. xiiThe subject of this work is concerned with the application of sliding mode control for indirect vector controlled induction machine drive systems. The motion of the control system employing variable control system can be described as having two modes: reaching and sliding modes. The reaching mode means the control mode before the states of the system reach the predefined sliding surface in which the robustness of the variable structure control technique is not guaranteed. Thus, the predefined control characteristics cannot be achieved. For the design of the speed control system, the induction machine is often modelled as a first-order system neglecting the electrical dynamics. Conventional variable structure control is not suitable for the first-order sys tem because the whole transient state belongs to the reaching mode and the robustness against the unknown disturbance is not guaranteed in the whole transient state. Because of this, most applications of variable structure control for the control of electric motors are limited to the position servo system having second-order dynamics. To overcome this problem, the acceleration of the motor is introduced. It is, however, difficult to obtain the information on the acceleration since the accel eration signal has generally high bandwidth and is very noisy. Chattering is an inherent feature of variable structure control and causes the problems of producing vibration and wearing the mechanical parts of the ac tuator. In particular, it is highly undesirable for the speed control system of the induction machine, because the induction machine can be operated at low speed as well as high speed and under different load conditions. Therefore, the reduction technique of chattering should be considered for designing the speed control system of the induction machine. Theoretically, variable structure control is based on the assumption of the infi nite switching action on the sliding surface. It is, however, difficult to meet this assumption in practice because of the limitations of the sampling frequency and control input. Thus, the ideal sliding motion having infinite switching action cannot be realized and this nonideal sliding motion is a factor of producing the undesirable steady-state error. In this scheme the sliding surface can be selected as follows. Define the error as where x represents one of the induction machine control variable which can be either the speed u>m, rotor flux ipr or stator currents ids and iqs. The sliding surface for the integral variable structure control can be given as /e(r)di -oo S = e + c I e{r)dT -oo xiiiwhere c is a positive constant. The control law can be designed as the sum of the equivalent control ueq and the switching control u3W. U = Ueq ~r Usw Unlike conventional variable structure control, the proposed integral variable structure control scheme provides not only a complete robustness without reach ing problem but also a good steady-state performance by the integral action. xiv
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